The need for a high-quality factor (Q), low insertion loss tunable filter pervades a wide range of microwave and RF applications, in both the military, e.g., RADAR, communications and Electronic Intelligence (ELINT), and the commercial fields such as in various communications applications, including cellular. Placing a sharply defined bandpass filter directly at the receiver antenna input will often eliminate various adverse effects resulting from strong interfeing signals at frequencies near the desired signal frequency in such applications. Because of the location of the filter at the receiver antenna input, the insertion loss must be very low to not degrade the noise figure. In most filter technologies, achieving a low insertion loss requires a corresponding compromise in filter steepness or selectivity. In the present invention, the extremely low loss property of high-temperature superconductor (HTS) filter elements provides an attractive solution, achieving a very low insertion loss yet simultaneously allowing a high selectivity/steepness bandpass definition.
In many applications, particularly where frequency hopping is used, a receiver filter must be tunable to either select a desired frequency or to trap an interfering signal frequency. The vast majority of lumped element tunable filters have used varactor diodes. Such a design amounts to using a tunable capacitor because varactor diodes, by changing the reverse bias voltage, vary the depletion thickness and hence the P-N junction capacitance. While varactors are simple and robust, they have limited Q's, and suffer from the problem that the linear process that tunes them extends all of the way to the signal frequency, so that high-amplitude signals create, through the resulting nonlinearities, undesirable intermodulation products and other problems.
Consider the case of a conventional varactor diode. In a varactor, the motion of electrons accomplishes the tuning itself. As the reverse bias voltage (Vr) on the junction of the varactor is changed, then in accordance with Poisson's Equation, the width of the P-N junction depletion region changes, which alters the junction capacitance (Cj). Because the tuning mechanism of varactors is electronic, the tuning speed is extremely fast. Unfortunately, this also leads to a serious associated disadvantage: limited dynamic range. Because the Cj (Vr) relationship is nearly instantaneous in response, extending to changes in Vr at the signal frequency itself, and the input signal (frequently in a resonantly magnified form) appears as a component of the junction bias voltage Vr, the input signal itself parametrically modulates the junction capacitance. If the signal amplitude across the varactor is very small in comparison to the dc bias, the effect is not too serious. Unfortunately, for high signal amplitudes, this parametric modulation of the capacitance can produce severe cross-modulation (IM) effects between different signals, as well as harmonic generation and other undesirable effects. While these signal-frequency varactor capacitance variations are the basis of useful devices such as parametric amplifiers, subharmonic oscillators, frequency multipliers, and many other useful microwave circuits, in the signal paths of conventional receivers they are an anathema. This inherent intermodulation or dynamic range problem will presumably extend to “tunable materials”, such as ferroelectrics or other materials in which the change of dielectric constant (εr) with applied electric field (E) is exploited to tune a circuit. As long as the εr (E) relationship applies out to the signal frequency, then the presence of the signal as a component of E will lead to the same intermodulation problems that the varactors have.
In addition to the intermodulation/dynamic range problems of varactors, these conventional tuning devices also have serious limitations in Q, or tuning selectivity. Because the varactors operate by varying the depletion region width of a P-N junction, at lower reverse bias voltages (higher capacitances), there is a substantial amount of undepleted moderately-doped semiconductor material between the contacts and the P-N junction that offers significant series resistance (Rac) to ac current flow. Since the Q of a varactor having junction capacitance Cj and series resistance Rac at an input signal frequency f is given by Q=1/(2 f Cj Rac), the varactor Q values are limited, particularly at higher frequencies. For example, a typical commercial varactor might have Cj=2.35 pF with Rac=1.0 Ω at Vr=−4V, or Cj=1.70 pF with Rac=0.82 Ωat Vr=−10V, corresponding to Q values at f=1.0 GHz of Q=68 at V, =4V or Q=114 at Vr=−10V (or f=10.0 GHz values of Q=6.8 and Q=11.4, respectively). Considering that an interesting X-band (f=10 GHz) RADAR application might want a bandwidth of Δf=20 MHz (FWHM), corresponding to a Q=f/Δf=500 quality factor, we see that available varactors have inadequate Q (too much loss) to meet such requirements. While the mechanisms are different, this will very likely apply to the use of ferroelectrics or other “tunable materials.” A general characteristic of materials which exhibit the field-dependent dielectric constant nonlinearities (that makes them tunable) is that they exhibit substantial values of the imaginary part of the dielectric constant (or equivalently, loss tangent). This makes it unlikely that, as in varactors, these “tunable materials” will be capable of achieving high Q's, particularly at high signal frequencies.
An additional problem with both varactors and “tunable materials” for circuits with high values of Q is that these are basically two-terminal devices; that is, the dc tuning voltage must be applied between the same two electrodes to which the signal voltage is applied. The standard technique is to apply the dc tuning bias through a “bias tee”-like circuit designed to represents a high reactive impedance to the signal frequency to prevent loss of signal power out the bias port (as this., loss would effectively reduce the Q). However, while the design of bias circuits that limit the loss of energy to a percent, or a fraction of a percent of the resonator energy is not difficult, even losses of a fraction of a percent are not nearly good enough for very high Q circuits (e.g., Q's in the 103 to >105 range, as achievable with HTS resonators). It would be much easier to design such very high Q circuits using three-terminal, or preferably 4-terminal (two-port) variable capacitors in which the tuning voltage is applied to a completely different pair of electrodes from those across which the input signal voltage is applied (with an inherent high degree of isolation between the signal and bias ports).
One new form of variable capacitor that avoids the intermodulation/dynamic range problems of varactors or “tunable materials” approaches is the microelectromechanical (HEMS) variable capacitor. A number of MEMS variable capacitor device structures have been proposed, including elaborate lateral-motion interdigitated electrode capacitor structures. In the simple vertical motion, parallel plate form of this device, a thin layer of dielectric separating normal metal plates (or a normal metal plate from very heavily doped silicon) is etched out in processing to leave a very narrow gap between the plates. The thin top plate is suspended on four highly compliant thin beams which terminate on posts (regions under which the spacer dielectric has not been removed). The device is ordinarily operated in an evacuated package to allow substantial voltages to be applied across the narrow gap between plates without air breakdown (and to eliminate air effects on the motion of the plate and noise). When a dc tuning voltage is applied between the plates, the small electrostatic attractive force, due to the high compliance of the support beams, causes substantial deflection of the movable plate toward the fixed plate or substrate, increasing the capacitance.
Because the change of capacitance, at least in the metal-to-metal plate version of the MEMS variable capacitor, is due entirely to mechanical motion of the plate (as opposed to “instantaneous” electronic motion effects as in varactors or “tunable materials”), the frequency response is limited by the plate mass to far below signal frequencies of interest. Consequently, these MEMS devices will be free of measurable intermodulation or harmonic distortion effects, or other dynamic range problems (up to the point where the combination of bias plus signal voltage across the narrow gap between plates begins to lead to nonlinear current leakage or breakdown effects).
In addition to their freedom from intermodulation/dynamic range problems, normal metal plate MEMS variable capacitor structures offer the potential for substantially lower losses and higher Q's. While the simple parallel plate MEMS structure has a Q problem due to the skin effect resistance, Rac, of the long narrow metal leads down the compliant beams supporting the movable plate, an alternative structure is possible which avoids this problem. If the top (movable) plate is made electrically “floating” (from a signal standpoint, it would still have a dc bias lead on it), and the fixed bottom plate split into two equal parts, these two split plates can be used as the signal leads to the MEMS variable capacitor. (The capacitance value is halved, of course, but the tuning range is preserved.) In this “floating plate” configuration, passage of ac current through the long narrow beam leads is avoided, allowing fairly high values of Q to be achieved, even with normal metal plates.
While this conventional MEMS variable capacitor structure is capable of improved Q's and avoids the intermodulation problems of varactors and “tunable materials”, it has some potential problems of its own. For example, the electrostatic force attracting the two plates is quite weak, except at extremely short range. The electrostatic force Fe between two parallel plates each of area A with a voltage difference V and a gap separation z is given byFe=−(ε0A/2)(V/z)2  (Eq. 1) where ε=8.854×10−12 Farad/Meter (F/m) is the permittivity of a vacuum. The extremely rapid falloff of force as the separation gap is increased (as 1/z2) makes the useful tuning range of electrostatic drivers quite small. In this parallel-plate MEMS capacitor configuration, if a linear spring provides the restoring force between the plates, when the bias voltage is increased such that the gap separation has dropped to ⅓ of the separation at zero bias, the plate motion becomes unstable and the plates snap together. This limits the useful tuning range to less than 3:1 in capacitance, or less than 1.732:1 in frequency. Further, the short-range nature of the electrostatic force makes its use in variable-inductance tuning even more problematic because of the requirement for very narrow gaps (to give reasonable levels of force at reasonable drive voltages), since much larger gaps (e.g., hundreds of microns) are desirable in devices having such variable-inductance tuning.
The short-range nature of the electrostatic force is illustrated by the following example. In a parallel-plate capacitor having a voltage of 100 volts (which is actually an unreasonably high voltage level given the trends toward low voltage electronics) and a gap separation of 1.0μ meter (μm), the electrostatic force (divided by the area of the plates) is 4.514 grams/centimeter2, a reasonable force. Increasing the gap to 10 μm at the same voltage produces the minuscule attractive force of 0.04514 grams/centimeter2. On the other hand, decreasing the gap to 0.1 μm at the same voltage produces the robust attractive force of 451.43 gramms/centimeter2, corresponding to an electric field strength of 107 V/cm. Although coating the plates with a thin dielectric and allowing progressive contact of thin curved (stress-bent) layers with a fixed electrode as voltage is increased may counteract the short-range effect of this electrostatic force (and with proper drive plate shaping, extend the tuning range in capacitance beyond 3:1), triboelectric (i.e., charging due to friction) and charge transfer effects under the high field condition tend to give significant hysteresis in the capacitance-voltage (C-V) characteristics of these “window shade” MEMS devices.
In addition, there are other potential problems in conventional MEMS devices. For example, in many system applications for tunable filters, requirements for precise phase make it essential that the selected frequency be very stable and reproducible. Consider a resonator or narrowband filter having a center frequency Fo and a −3 dB bandwidth ΔF given from its (loaded) quality factor Qo by the equationΔF=Fo/Qo  (Eq. 2) Note that as the frequency is changed from (Fo−ΔF/2) through Fo to (Fo+ΔF/2), the phase changes quite dramatically from +45° to 0° to −45°. For a signal frequency f near Fo, the phase in a single resonator may be approximated byPhase(°)≈2Qo(180°/π)[1−(f/Fo)]  (Eq. 3) (for a single resonator, or Nr times this value for a filter having Nr resonators at Fo). Hence, if the allowable phase uncertainty at a given frequency f is denoted by ΔPhase (°), then the allowable error in the resonator center frequency, ΔFo, near resonance will beΔFo/f=ΔPhase (°)/[2 Qo(180°/π)]=(0.0087266/Qo)ΔPhase(°)  (Eq. 4) For example, for a 1.0° degree phase error with a loaded Qo=500, the resonator frequency repeatability, ΔFo/f, must be less than or equal to 0.00175% (for a single resonator, or 1/Nr times this value for a number Nr of resonators). This means that for such phase sensitive applications, the tunable elements must achieve levels of repeatability, hysteresis and continuity that appear difficult to achieve in ferroelectric piezoelectric actuators, let alone “window shade” electrostatic MEMS devices.
Therefore, there is a need in the art for new driver structures for varying the properties of MEMS-like HTS capacitors or inductors, or more complex distributed resonator structures having transmission line-like qualities. The resulting variable capacitors, inductors, or other tunable elements may be incorporated into tunable filters or other circuits.